Voltage regulator

ABSTRACT

A voltage regulator includes a current bridge and first and second current paths coupling a current mirror to respective first and second voltage-to-current converters. The current mirror controls a second current dependent on a first current. The first voltage-to-current converter controls the first current dependent on either a reference voltage or a feedback voltage derived from the regulator&#39;s output voltage, and the second voltage-to-current converter controls the second current dependent on the other of the feedback and reference voltages. Voltage-to-current conversion by the first converter is independent of voltage-to-current conversion by the second converter. An output transistor stage coupled to the second current path controls the output voltage dependent on the voltage in the second current path indicative of a deviation of the second current from a target current value dependent on the reference voltage.

This application is a continuation-in-part of International ApplicationNo. PCT/EP2011/055047 filed on Mar. 31, 2011, which claims the benefitof the filing date of U.S. Provisional Patent Application No. 61/325,887filed on Apr. 20, 2010, and which claims priority to European PatentApplication 10250718.3 filed on Apr. 1, 2010. Those three applicationsare incorporated in this application by reference.

TECHNICAL FIELD

The present disclosure relates to a voltage regulator and to a method ofregulating an output voltage, and has application in, particularly butnot exclusively, integrated circuits and power supply circuits forintegrated circuits.

BACKGROUND

Low drop-out (LDO) voltage regulators are widely used to supply power tointegrated circuits due to their ability to operate at a low voltage andtheir high power efficiency. An LDO voltage regulator is a voltageregulator which is able to regulate an output voltage to a predefinedvalue with a very low difference between an input voltage and the outputvoltage. Such a voltage regulator may be embedded in an integratedcircuit or may be provided externally.

A typical LDO voltage regulator known in the prior art comprises anoutput stage implemented as common source or common emitter transistoramplifier and an error amplifier arranged in a regulation loop whichgenerates an error signal by comparing the output voltage to a referencevoltage and which drives the output stage with the error signal.

An LDO voltage regulator 30 suitable for implementation in aComplementary Metal Oxide Semiconductor (CMOS) device is illustrated inFIG. 1. An input voltage V_(DD) is supplied to a source of an outputtransistor 14, which is a p-channel metal oxide semiconductor fieldeffect transistor (MOSFET), and the output voltage V_(OUT) is deliveredat a drain of the output transistor 14. Coupled between the drain of theoutput transistor 14 and a node, which may be a ground, are seriescoupled resistors R₁ and R₂. The junction of the series coupledresistors R₁, R₂ is coupled to a non-inverting input of an erroramplifier 12. An inverting input of the error amplifier 12 is coupled toa reference voltage V_(REF), and an output of the error amplifier 12 iscoupled to a gate of the output transistor 14. The output voltageV_(OUT) is delivered to a load, which is represented by a load resistiveelement R_(L) coupled to the drain of the output transistor 14. In orderto decouple the voltage regulator 30 from the load, a load capacitiveelement C_(L) is coupled to the drain of the output transistor 14 inparallel with the load resistive element R_(L). In order to ensurestability, a series coupled feedback capacitor C_(F) and feedbackresistor R_(F) are coupled between the drain and a gate of the outputtransistor 14. The feedback capacitor C_(F) can require a large siliconarea for implementation in an integrated circuit. The load capacitiveelement C_(L) can require an even larger silicon area, or cannecessitate the use of an external discrete component. The use of anexternal discrete component can be undesirable due to the additionalspace required and parasitic components introduced by additionalinterconnections. Furthermore, the presence of the feedback capacitorC_(F) can reduce the speed of operation of the voltage regulator 30,resulting in fast changes in the output voltage V_(OUT) when fastchanges occur in the current drawn by a load coupled to the outputvoltage V_(OUT), such as can occur when parts of load circuits areswitched on and off for power conservation. Fast changes in the outputvoltage V_(OUT) can be reduced by means of filtering using a suitablylarge load capacitive element C_(L), although the load capacitiveelement C_(L) can also reduce the stability of the voltage regulator 30,which can oscillate if the load capacitive element C_(L) is very large.

An alternative voltage regulator 40 known in the prior art isillustrated in FIG. 2. Its architecture differs from the architecture ofthe LDO voltage regulator 30 of FIG. 1 in two respects. First, itsoutput stage comprises an n-channel MOSFET output transistor 16 with itsdrain coupled to the input voltage V_(DD) and the output voltage V_(OUT)delivered at its source. This configuration has improved stability,because the output transistor 16 normally doesn't introduce a dominantpole in the frequency range where the voltage regulator 40 has gain.Second, due to the improved stability, the feedback capacitor C_(F) andfeedback resistor R_(F) of the LDO voltage regulator of FIG. 1 areomitted. However, the voltage regulator 40 of FIG. 2 is not an LDOvoltage regulator. This is because the error amplifier 12 has to becapable of delivering at its output a voltage exceeding V_(OUT)+V_(GS),where V_(GS) is the gate-source threshold voltage of the outputtransistor 16 which is normally in the range 0.6 to 0.7 volts, andtherefore the input voltage V_(DD) must also exceed V_(OUT)+V_(GS).

A further voltage regulator 50 known in the prior art is illustrated inFIG. 3. Its architecture differs from the architecture of the voltageregulator 40 of FIG. 2 by employing a charge pump 18 to convert theinput voltage V_(DD) to a higher voltage V_(H), for example double theoutput voltage V_(DD), by charging a storage capacitor C_(Q2). Thehigher voltage V_(H) is supplied to the error amplifier 12. Thisarchitecture can enable LDO operation. However, the storage capacitorC_(Q2), and a pump capacitor C_(Q1) required for the operation of thecharge pump 18, can require a large silicon area for implementation inan integrated circuit, and the higher voltage V_(H) may exceed thetechnological limits of modern sub-micron technologies. Also, thisarchitecture can result in increased power consumption.

SUMMARY

According to a first aspect, there is provided a voltage regulatorcomprising:

a first input for a first input voltage;

a second input for a second input voltage lower than the first inputvoltage;

an output for an output voltage;

an output transistor stage having a first terminal coupled to a firstone of the first and second inputs, a second terminal coupled to theoutput, and a control terminal for controlling the conductivity of theoutput transistor stage between the first terminal and the secondterminal;

a feedback network coupled between the output and a second one of thefirst and second inputs, being different from the first one of the firstand second inputs, and arranged to produce at a feedback node a feedbackvoltage dependent on the output voltage;

a first current path for conveying a first current and a second currentpath for conveying a second current;

a primary current mirror stage coupled to the first current path and tothe second current path and arranged to control the second currentdependent on the first current;

a first voltage-to-current converter coupled to the first current pathand arranged to control the first current dependent on one of thefeedback voltage and a reference voltage, and a secondvoltage-to-current converter coupled to the second current path andarranged to control the second current dependent on the other of thefeedback voltage and the reference voltage, wherein thevoltage-to-current conversion provided by the first voltage-to-currentconverter is independent of the voltage-to-current conversion providedby the second voltage-to-current converter; wherein the control terminalis coupled to the second current path for controlling the conductivityof the output transistor stage dependent on a voltage in the secondcurrent path indicative of a deviation of the second current from atarget current value dependent on the reference voltage for therebyreducing a deviation of the output voltage from a target voltage value.

According to a second aspect, there is provided a method of regulatingan output voltage, the method comprising:

producing a feedback voltage dependent on the output voltage;

controlling a first current in a first current path dependent on one ofthe feedback voltage and a reference voltage by means of a firstvoltage-to-current converter;

controlling a second current in a second current path dependent on thefirst current by means of a primary current mirror stage and controllingthe second current dependent on the other of the feedback voltage andthe reference voltage by means of a second voltage-to-current converter,wherein the voltage-to-current conversion provided by the firstvoltage-to-current converter is independent of the voltage-to-currentconversion provided by the second voltage-to-current converter; and

reducing a deviation of the output voltage from a target voltage valueby controlling the output voltage dependent on a voltage in the secondcurrent path indicative of a deviation of the second current from atarget current value dependent on the reference voltage.

The first current path and the second current path may be considered tobe branches of a bridge circuit, with the current in one current pathbeing dependent on the feedback voltage, and the current in the othercurrent path being dependent on the reference voltage. Also, by means ofthe primary current mirror stage, the current in one path is areflection of the current in the other path. The bridge will be balancedwhen the currents in the first and second current paths are matched,according to a current mirror ratio of the primary current mirror stage.The output voltage is controlled dependent on a voltage in the secondcurrent path, and will be at a target value when the bridge is balanced.

The voltage regulator according to the first aspect and the method ofregulating an output voltage according to the second aspect areadvantageous in the following respects:

-   -   LDO operation or non-LDO operation can be provided;    -   fast operation is enabled;    -   stable operation is enabled with a wide range of load current        and load capacitance;    -   the load capacitive element C_(L) can be dispensed with, or can        be of reduced size;    -   the feedback capacitor C_(F) and feedback resistor R_(F) of the        prior art illustrated in FIG. 1 can be dispensed with, enabling        a stable voltage regulator to be implemented without capacitors,        or they can be of reduced size;    -   the use of the charge pump 18, the pump capacitor C_(Q1) and the        storage capacitor C_(Q2) of the prior art illustrated in FIG. 3        can be avoided; and    -   a positive or negative output voltage can be provided.

Optionally, the first voltage-to-current converter can comprise a firsttransconductance amplifier having a first transconductance amplifierfirst input coupled to the second one of the first and second inputs viaa first current sensing resistive element, a first transconductanceamplifier second input arranged to receive the one of the feedbackvoltage and the reference voltage, and a first transconductanceamplifier output coupled to control the conductivity of a first currentconverter transistor dependent on a difference between a voltage at thefirst transconductance amplifier first input and a voltage at the firsttransconductance amplifier second input, wherein the first currentconverter transistor is arranged to control the first current in thefirst current path, and the second voltage-to-current converter cancomprise a second transconductance amplifier having a secondtransconductance amplifier first input coupled to the second one of thefirst and second inputs via a second current sensing resistive element,a second transconductance amplifier second input arranged to receive theother of the feedback voltage and the reference voltage, and a secondtransconductance amplifier output coupled to control the conductivity ofa second current converter transistor dependent on a difference betweena voltage at the second transconductance amplifier first input and avoltage at the second transconductance amplifier second input, whereinthe second current converter transistor is arranged to control thesecond current in the second current path. Such voltage-to-currentconverters can enable fast operation of the voltage regulator.

Optionally, the one of the first and second inputs can be the firstinput and the other of the first and second inputs can be the secondinput, and the output transistor stage can comprise an output transistorhaving a p-channel, a source coupled to the first terminal, a draincoupled to the second terminal and a gate coupled to the controlterminal. This embodiment enables LDO operation of the voltage regulatorfor a positive output voltage.

Optionally, the one of the first and second inputs can be the firstinput and the other of the first and second inputs can be the secondinput, and the output transistor stage can comprise an output transistorhaving an n-channel, a drain coupled to the first terminal, a sourcecoupled to the second terminal and a gate coupled to the controlterminal. This embodiment enables non-LDO operation of the voltageregulator for a positive output voltage.

Optionally, the one of the first and second inputs can be the secondinput and the other of the first and second inputs can be the firstinput, and the output transistor stage can comprise an output transistorhaving an n-channel, a source coupled to the first terminal, a draincoupled to the second terminal and a gate coupled to the controlterminal. This embodiment enables LDO operation of the voltage regulatorfor a negative output voltage.

Optionally, the one of the first and second inputs can be the secondinput and the other of the first and second inputs can be the firstinput, and the output transistor stage can comprise an output transistorhaving a p-channel, a drain coupled to the first terminal, a sourcecoupled to the second terminal and a gate coupled to the controlterminal. This embodiment enables non-LDO operation of the voltageregulator for a negative output voltage.

Optionally, the first and second current converter transistors can eachcomprise an n-channel, the first transconductance amplifier first inputand the second transconductance amplifier first input can be invertinginputs, and the first transconductance amplifier second input and thesecond transconductance amplifier second input can be non-invertinginputs. This embodiment enables regulation of a positive output voltageusing n-channel transistors in the first and second voltage-to-currentconverters.

Optionally, the first and second current converter transistors can eachcomprise a p-channel, the first transconductance amplifier first inputand the second transconductance amplifier first input can be invertinginputs, and the first transconductance amplifier second input and thesecond transconductance amplifier second input can be non-invertinginputs. This embodiment enables regulation of a negative output voltageusing p-channel transistors in the first and second voltage-to-currentconverters.

Optionally, the first current sensing resistive element and the firstcurrent converter transistor can be arranged in the first current pathand the second current sensing resistive element and the second currentconverter transistor can be arranged in the second current path. Thisembodiment enables a simple implementation.

Optionally, a first secondary current mirror stage can be coupledbetween the first current path and the first voltage-to-currentconverter for controlling the first current dependent on a reflection ofa current in the first voltage-to-current converter, and a secondsecondary current mirror stage can be coupled between the second currentpath and the second voltage-to-current converter for controlling thesecond current dependent on a reflection of a current in the secondvoltage-to-current converter. Likewise, the method can comprisecontrolling the first current dependent on a reflection of a current inthe first voltage-to-current converter, and controlling the secondcurrent dependent on a reflection of a current in the secondvoltage-to-current converter. This feature can provide a versatilearchitecture which enables the voltage regulator to be implemented usinga plurality of identical cells according to the magnitude of a requiredoutput current.

Optionally, the first current path can comprise a plurality of firstcurrent sub-paths for each conveying a proportion of the first current,the second current path can comprise a plurality of second currentsub-paths for each conveying a proportion of the second current, theprimary current mirror stage can comprise a plurality of primary currentmirror devices, the first secondary current mirror stage can comprise aplurality of first secondary current mirror devices coupled torespective ones of the primary current mirror devices by means of therespective first current sub-paths, the second secondary current mirrorstage can comprise a plurality of second secondary current mirrordevices coupled to respective ones of the primary current mirror devicesby means of the respective second current sub-paths, and the outputtransistor stage can comprise a plurality of output transistors coupledbetween the first one of the first and second inputs and the output,wherein each of the output transistors is coupled to a different one ofthe second current sub-paths for controlling the conductivity of therespective output transistor between the first one of the first andsecond inputs and the output dependent on a voltage in the respectivesecond current sub-path. Likewise, the method optionally can compriseconveying a proportion of the first current via each of a plurality offirst current sub-paths and conveying a proportion of the second currentvia each of a plurality of second current sub-paths, and controlling,dependent on a voltage in the respective current sub-path, theconductivity of each of a plurality of output transistors coupled to adifferent one of the first or second current sub-paths. This feature canprovide a versatile architecture which enables the voltage regulator tobe implemented using a plurality of identical cells according to themagnitude of a required output current.

Optionally, the primary current mirror stage can be arranged to controlthe second current to be equal to the first current. Likewise, themethod optionally can comprise controlling the second current to beequal to the first current. This feature can enable close matching ofthe first and second currents and also improved speed and stability.

Optionally, the primary current mirror stage can be arranged to controlthe second current to be greater than the first current. Likewise, themethod optionally can comprise controlling the second current to begreater than the first current. This feature can enable powerconsumption of the voltage regulator to be reduced.

Optionally, the voltage regulator can comprise a differential amplifierstage coupled to the primary current mirror stage by means of a thirdcurrent path for conveying a third current and by means of a fourthcurrent path for conveying a fourth current, and coupled to the feedbacknetwork for receiving the feedback voltage, wherein the differentialamplifier stage is arranged to control the third current dependent onthe one of the feedback voltage and the reference voltage and to controlthe fourth current dependent on the other of the feedback voltage andthe reference voltage, and wherein the primary current mirror stage isarranged to control the fourth current dependent on the third current.Likewise, the method optionally can comprise conveying a third currentbetween a differential amplifier stage and the primary current mirrorstage by means of a third current path, conveying a fourth currentbetween the differential amplifier stage and the primary current mirrorstage by means of a fourth current path, employing the differentialamplifier stage to control the third current dependent on one of thefeedback voltage the reference voltage and to control the fourth currentdependent on the other of the feedback voltage and the referencevoltage, and employing the primary current mirror stage to control thefourth current dependent on the third current. This feature can enablethe voltage regulator to have a higher gain and bandwidth.

Optionally, the differential amplifier is arranged to control the thirdcurrent to be smaller than the first current and the fourth current tobe smaller than the second current by, for example, a factor of at leastten. This feature can contribute to the voltage regulator having a highstability and high phase margin.

Optionally, the voltage regulator can comprise a capacitive elementcoupled between the output and the feedback node. This feature canenable fast operation of the voltage regulator.

Optionally, the voltage regulator can comprise a capacitive elementcoupled between the output and one of the first and second inputs. Thisfeature can decouple the voltage regulator from a load coupled to theoutput.

Optionally, the voltage regulator can be formed in an integratedcircuit.

According to a further aspect there is provided an electronic apparatuscomprising a voltage regulator according to the first aspect.

BRIEF DESCRIPTION OF THE DRAWINGS

Preferred embodiments will now be described, by way of example only,with reference to the accompanying drawings, in which:

FIG. 1 is a schematic diagram of a prior art voltage regulator;

FIG. 2 is a schematic diagram of a prior art voltage regulator;

FIG. 3 is a schematic diagram of a prior art voltage regulator;

FIG. 4 is a schematic diagram of a voltage regulator in accordance withan embodiment of the invention;

FIG. 5 is a schematic diagram of voltage-to-current converters;

FIG. 6 is a schematic diagram of a primary current mirror stage;

FIG. 7 is a schematic diagram of a voltage regulator for a positivevoltage and LDO operation;

FIG. 8 is a schematic diagram of a voltage regulator for a negativevoltage and LDO operation;

FIG. 9 is a schematic diagram of a voltage regulator for a positivevoltage and non-LDO operation;

FIG. 10 is a schematic diagram of a voltage regulator for a negativevoltage and non-LDO operation;

FIG. 11 is a schematic diagram of a voltage regulator for a positivevoltage and including a differential amplifier;

FIG. 12 is a schematic diagram of a voltage regulator for a negativevoltage and including a differential amplifier;

FIG. 13 is a schematic diagram of a primary current mirror stage;

FIG. 14 is a schematic diagram of a voltage regulator with additionalcurrent mirroring;

FIG. 15 is a schematic diagram of a voltage regulator with a modularstructure;

FIG. 16 is a schematic diagram of an electronic apparatus comprising avoltage regulator;

FIG. 17 is a schematic diagram of a voltage regulator for a plurality ofpositive output voltages;

FIG. 18 is a schematic diagram of a voltage regulator for a plurality ofnegative output voltages; and

FIG. 19 is a schematic diagram of a voltage regulator with additionalcurrent mirroring and a plurality of output voltages.

DETAILED DESCRIPTION

Referring to FIG. 4, a voltage regulator 100 comprises a first input 102for a first input voltage V_(IN1), a second input 106 for a second inputvoltage V_(IN2) lower than the first input voltage V_(IN1), which may bea ground, and an output 104 for an output voltage V_(OUT). An outputtransistor stage 110 has a first terminal 112 coupled to the input 102,a second terminal 114 coupled to the output 104, and a control terminal116 for controlling the conductivity of the output transistor stage 110between the first terminal 112 and the second terminal 114. The outputtransistor stage 110 illustrated in FIG. 4 comprises a p-channel outputtransistor MP which is a p-channel MOSFET in a common sourceconfiguration, having a source coupled to the first terminal 112, adrain coupled to the second terminal 114 and a gate coupled to thecontrol terminal 116. This configuration can provide LDO operation.

Coupled to the output 104 of the voltage regulator 100 is a feedbacknetwork 120 arranged to produce a feedback voltage V_(FB) dependent onthe output voltage V_(OUT). The feedback network 120 illustrated in FIG.4 comprises feedback resistors R₁, R₂ coupled in series between theoutput 104 and the second input 106, thereby forming a voltage divider,although other arrangements of the feedback network 120 may be used. Ajunction between the feedback resistors R₁, R₂ is coupled to a feedbacknode 108 for delivering the feedback voltage V_(FB). Coupled between theoutput 104 of the voltage regulator 100 and the feedback node 108 atwhich the feedback voltage V_(FB) is delivered is an optional feedbackcapacitive element C_(B), which can facilitate fast operation of thevoltage regulator 100 by increasing gain at high frequencies.

The voltage regulator 100 comprises a first current path 160 forconveying a first current I1 and a second current path 162 for conveyinga second current I2. There is a primary current mirror stage 130 coupledto the first current path 160 and to the second current path 162, andthe primary current mirror stage 130 is arranged to control the secondcurrent I2 dependent on the first current I1 by mirroring the firstcurrent I1 such that the second current I2 is a reflection, or mirror,of the first current I1. More specifically, the second current I2 isrelated to the first current I1 by a current mirror ratio M, that is,I2=M·I1. The second current I2 may be controlled to be equal to thefirst current I1, in which case the value of the current mirror ratio Mis one, or alternatively the second current I2 may be controlled to begreater than the first current I1, in which case the value of thecurrent mirror ratio M is greater than one. The primary current mirrorstage 130 is coupled to the first input 102 of the voltage regulator 100for deriving power from the first input voltage V_(IN1), althoughalternatively the primary current mirror stage 130 may be powered from adifferent supply.

A first voltage-to-current converter 150 is coupled to the first currentpath 160 and to the feedback node 108, and is arranged to control thefirst current I1 dependent on the feedback voltage V_(FB). The firstvoltage-to-current converter 150 is also arranged to receive the secondinput voltage V_(IN2) applied at the second input 106 by means of afirst connection 168. The first connection 168 conveys the first currentI1 controlled by the first voltage-to-current converter 150. A secondvoltage-to-current converter 155 is coupled to the second current path162 and to a reference voltage V_(REF), and is arranged to control thesecond current I2 dependent on the reference voltage V_(REF). Thereference voltage V_(REF) can be provided by, for example, a band-gapdevice. The second voltage-to-current converter 155 is arranged toreceive the second input voltage V_(IN2) by means of a second connection170. The second connection conveys the second current I2 controlled bythe second voltage-to-current converter 155. The first and secondconnections 168, 170 are separate, that is they provide independentcurrent paths. This enables the voltage-to-current conversion performedby the second voltage-to-current converter 155 to be independent of thevoltage-to-current conversion performed by the first voltage-to-currentconverter 150. Nevertheless, because changes to the first current I1resulting from changes in the feedback voltage V_(FB) are reflected inthe second current I2 by the primary current mirror stage 130, thecontrol of the second current I2 due to the reference voltage V_(REF)can be linearly superimposed on the changes in second current I2 due tothe changes in the feedback voltage V_(FB).

The control terminal 116 of the output transistor stage 110 is coupledto the second current path 162 for controlling the conductivity of theoutput transistor stage 110 between the first terminal 112 and thesecond terminal 114 dependent on a voltage in the second current path162.

In operation, the primary current mirror stage 130, the first and secondvoltage-to-current converters 150, 155 and the first and second currentpaths 160, 162 form a current bridge. The bridge is balanced when theratio of the second current I2 to the first current I1 is equal, orclose, to the current mirror ratio M, and in this state the voltage inthe first current path 160 between the primary current mirror stage 130and the first voltage-to-current converter 150, and the voltage in thesecond current path 162 between the primary current mirror stage 130 andthe second voltage-to-current converter 155, are equal, or similar. Alsowhen the bridge is balanced, the second current I2 is at a targetcurrent value determined by the reference voltage V_(REF), and theoutput voltage V_(OUT) is stable at a target voltage value dependent onthe reference voltage V_(REF). If the output voltage V_(OUT) deviatesfrom the target voltage value, for example if an additional load beginsto draw current from the output 104 of the voltage regulator 100, or adecreased load reduces the current drawn from the output 104 of thevoltage regulator 100, the feedback voltage V_(FB) will change. Inresponse to the change in the feedback voltage V_(FB), the firstvoltage-to-current converter 150 will operate to change the firstcurrent I1, thereby causing the current bridge to become unbalanced,meaning the ratio of the second current I2 to the first current I1 is nolonger equal, or close, to the current mirror ratio M, and that thevoltage in the first and second current paths 160, 162 is no longerequal, or similar. In response to the change in the first current I1,the primary current mirror stage 130 will operate to change the secondcurrent I2 to maintain the current mirror ratio M, and balance will berestored in the current bridge. For example, if the output voltageV_(OUT) increases above the target voltage value, then the feedbackvoltage V_(FB) will also increase, thereby causing the first current I1to increase and the voltage in the first current path 160 to decrease.In response, the second current I2 will increase and the voltage in thesecond current path 162 will increase. Preferably the secondvoltage-to-current converter 155 has a high output resistance, therebycausing the second current I2 to change very little from the targetcurrent value determined by the reference voltage V_(REF) despite alarge change in the voltage in the second current path 162. In thiscase, when the primary current mirror stage 130 operates to increase ordecrease the second current I2 by a small amount in response to a changein the first current I1, the voltage in second current path 162 willincrease or decrease by a larger amount. In response to the increase inthe voltage in the second current path 162, the voltage applied to thecontrol terminal 116 of the output transistor stage 110 will increase,thereby decreasing the voltage between the gate and the source of theoutput transistor MP, and thereby decreasing the conductivity of theoutput transistor stage 110 and resulting in a decrease in the outputvoltage V_(OUT). Alternatively, if the output voltage V_(Out) decreasesbelow the target value, then the feedback voltage V_(FB) will alsodecrease, thereby causing the first current I1 to decrease and thevoltage in the first current path 160 to increase. In response, thesecond current I2 will decrease and the voltage in the second currentpath 162 will decrease. In response to the decrease in the voltage inthe second current path 162, the voltage applied to the control terminal116 of the output transistor stage 110 will decrease, and the voltagebetween the gate and the source of the p-channel output transistor MPwill increase, thereby increasing the conductivity of the outputtransistor stage 110, resulting in an increase in the output voltageV_(OUT).

An embodiment of the first voltage-to-current converter 150 and thesecond voltage-to-current converter 155 is illustrated in FIG. 5.Referring to FIG. 5, the first voltage-to-current converter 150 has aninput for receiving the feedback voltage V_(FB) from the feedbacknetwork 120, an input for coupling to the first current path 160 forreceiving the first current I1, and an input for coupling to the secondinput 106 via the first connection 168 for receiving the second inputvoltage V_(IN2). The first voltage-to-current converter 150 comprises afirst transconductance amplifier T1 having a first inverting input 152coupled to the second input 106 via a first current sensing resistorR_(S1), a first non-inverting input 153 for coupling to the feedbacknode 108 for receiving the feedback voltage V_(FB), and a first output154 coupled to a first current converter transistor MN1 for controllingthe conductivity of the first current converter transistor MN1. Thefirst current converter transistor MN1 is coupled between the firstcurrent path 160 and the first current sensing resistor R_(S1). Thefirst current I1 passes through the first current converter transistorMN1, the first current sensing resistor R_(S1), and the first connection168.

Continuing to refer to FIG. 5, the second voltage-to-current converter155 has an input for receiving the reference voltage V_(REF), an inputfor coupling to the second current path 162 for receiving the secondcurrent I2, and an input for coupling to the second input 106 via thesecond connection 170 for receiving the second input voltage V_(IN2).The second voltage-to-current converter 155 comprises a secondtransconductance amplifier T2 having a second inverting input 156coupled to the second input 106 via a second current sensing resistorR_(S2), a second non-inverting input 157 for receiving the referencevoltage V_(REF), and a second output 158 coupled to a second currentconverter transistor MN2 for controlling the conductivity of the secondcurrent converter transistor MN2. The second current convertertransistor MN2 is coupled between the second current path 162 and thesecond current sensing resistor R_(S2). The second current I2 passesthrough the second current converter transistor MN2, the second currentsensing resistor R_(S2), and the second connection 170.

The first and second current converter transistors MN1, MN2 aren-channel metal oxide semiconductor (NMOS) transistors. The first andsecond transconductance amplifiers T1, T2 can each comprise a singlestage amplifier, such as a differential amplifier with or without afolded cascode or another configuration implementing a differentialinput. Power supply connections to the first and second transconductanceamplifiers T1, T2 are omitted from FIG. 5 for clarity.

In operation, first transconductance amplifier T1 compares the voltageon the first current sensing resistor R_(S1), which is applied to thefirst inverting input 152 of the first transconductance amplifier T1,with the feedback voltage V_(FB) applied to the first non-invertinginput 153 of the first transconductance amplifier T1, and the voltage atthe first output 154 of the first transconductance amplifier T1resulting from the comparison is applied to a gate of the first currentconverter transistor MN1. In this way, the first transconductanceamplifier T1 operates to align the voltage on the first current sensingresistor R_(S1) with the feedback voltage V_(FB), and in doing socontrols the first current I1 which flows through the first currentconverter transistor MN1 and the first current sensing resistor R_(S1).

The second transconductance amplifier T2 operates in a correspondingmanner, comparing the voltage on the second current sensing resistorR_(S2), which is applied to the second inverting input 156 of the secondtransconductance amplifier T2, with the reference voltage V_(REF)applied to the second non-inverting input 157 of the secondtransconductance amplifier T2. The voltage at the second output 158 ofthe second transconductance amplifier T2 resulting from the comparisonis applied to a gate of the second current converter transistor MN2. Inthis way, the second transconductance amplifier T2 operates to align thevoltage on the second current sensing resistor R_(S2) with the referencevoltage V_(REF), and in doing so controls the second current I2 whichflows through the second current converter transistor MN2 and the secondcurrent sensing resistor R_(S2). In this way, the firstvoltage-to-current converter 150 controls the first current I1 dependenton the feedback voltage V_(FB), and the second voltage-to-currentconverter 155 controls the second current I2 dependent on the referencevoltage V_(REF). In particular, the voltage at the junction of the firstcurrent sensing resistor R_(S1) and the first current convertertransistor MN1, which is applied to the first transconductance amplifierT1, and the voltage at the junction of the second current sensingresistor R_(S2) and the second current converter transistor MN2, whichis applied to the second transconductance amplifier T2 can be differentand can vary independently of each other. Other embodiments of the firstvoltage-to-current converter 150 and the second voltage-to-currentconverter 155 may alternatively be used.

Preferably the first and second current sensing resistors R_(S1) andR_(S2) are matched by being constructed using the same structure, forexample poly-silicon pieces with the same size, and by locating themclose to each other with the same orientation, although they need nothave equal values of resistance. This can enable the first and secondcurrent sensing resistors R_(S1) and R_(S2) to have proportionalresistance values and the same temperature dependence. In this way, anyinaccuracy in the resistance values can be of the same proportion and inthe same direction, thereby affecting both the first and second currentsI1 and I2 in the same way. If any input voltage offset introduced by thefirst and second transconductance amplifiers T1, T2 is neglected, thenthe first current I1 can be expressed as I1=(V_(OUT)·R2)/((R1+R2)·Rs1),where R1, R2 and R_(S1) represent, respectively the resistance of thefeedback resistors R1, R2 and the first current sensing resistor R_(S1),and the second current I2 can be expressed as I2=V_(REF)/R_(S2), whereR_(S2) represents the resistance of the second current sensing resistorR_(S2). If the bridge formed by the primary current mirror stage 130,the current control stage 140 and the first and second current paths160, 162 is balanced, then the output voltage V_(OUT) is equal to thetarget voltage value and can be expressed asV_(OUT)=V_(REF)·(R1+R2)·R_(S1)/M·R2·R_(S2), where M=I2/I1. If thecurrent mirror ratio M is one, resulting in the first and secondcurrents I1, I2 being equal, and if the first and second current sensingresistors R_(S1), R_(S2) are equal, then the target value of thefeedback voltage V_(FB) is equal to V_(REF) and so the target value ofthe output voltage V_(OUT) can be expressed asV_(OUT)=V_(REF)·(R1+R2)/R2.

In the voltage regulator 100 illustrated in FIG. 4, the first currentpath 160 drives only the first voltage-to-current converter 150. Incontrast, the second current path 162 drives the gate of the p-channeloutput transistor MP of the output transistor stage 110, in addition todelivering the second current I2 to the second voltage-to-currentconverter 155. Depending on the current to be drawn from the output 104of the voltage regulator 100, the p-channel output transistor MP may beof such a size that it presents a significant capacitive load to thesecond current path 162. In this case, the second current I2 in thesecond current path 162 may need to have a high value in order for thevoltage regulator 100 to operate at a sufficiently high speed.Therefore, in order to minimise power consumption, the first current I1may be arranged to have a lower value than the second current I2, inwhich case the current mirror ratio M is greater than one.

An embodiment of the primary current mirror stage 130 is illustrated inFIG. 6, and comprises a first current mirror transistor MP1 and a secondcurrent mirror transistor MP2, these both being p-channel metal oxidesemiconductor (PMOS) transistors. The first and second current mirrortransistors MP1, MP2 have their sources coupled to the first input 102for receiving the first input voltage V_(IN1) and their gates coupledtogether, thereby establishing common operating conditions for the firstand second current mirror transistors MP1, MP2. The first current mirrortransistor MP1 has its drain coupled to the first current path 160 fordelivering the first current I1, and its drain coupled to its gate forcontrolling the gate of both the first and second current mirrortransistors MP1, MP2 with a common voltage. The second current mirrortransistor MP2 has its drain coupled to the second current path 162 fordelivering the second current I2 reflected from the first current I1.For a current mirror ratio M of one, the first and second current mirrortransistors MP1, MP2 are of equal size, whereas for other values of thecurrent mirror ratio, the first and second current mirror transistorsMP1, MP2 can be of different sizes. Other embodiments of the primarycurrent mirror stage 130 may alternatively be used.

Further embodiments of voltage regulators are described below whichillustrate some of the variations that fall within the scope of theinvention, including the provision of a positive or a negative outputvoltage, the use of n-channel or p-channel transistors, the use of LDOor non-LDO operation, the use of the first and second currents I1, I2which flow either from the primary current mirror stage 130 to the firstand second voltage-to-current converters 150, 155 or in the oppositedirection, and the use of either the reference voltage V_(REF) or thefeedback voltage V_(FB) by either of the first and secondvoltage-to-current converters 150, 155 to control respectively the firstcurrent I1 and the second current I2. Despite the variations employed ineach of the embodiments of the voltage regulator, according to theterminology used throughout this description and the accompanyingclaims, for each embodiment the primary current mirror stage 130controls the second current I2 in the second current path 162 to be areflection of the first current I1 in the first current path 160, andthe control terminal 116 of the output transistor stage 110 is in eachembodiment coupled to the second current path 162 conveying the secondcurrent I2.

FIG. 7 illustrates a voltage regulator 200 having the same generalarchitecture as the voltage regulator 100 illustrated in FIG. 4 andincorporating the embodiments of the first and second voltage-to-currentconverters 150, 155 illustrated in FIG. 5 and the primary current mirrorstage 130 illustrated in FIG. 6. In FIG. 7 the optional feedbackcapacitive element C_(B) has been omitted. Furthermore in FIG. 7, andcorrespondingly in FIGS. 8 to 12, 14 and 15 illustrating furtherembodiments, a load resistive element R_(L) is coupled to the output 104and, although not part of the voltage regulator 200, illustrates how aload is coupled to the voltage regulator 200. In FIG. 7 the loadresistive element R_(L) is coupled between the output 104 and the secondinput 106. An optional load capacitive element C_(L) is coupled inparallel with the load resistive element R_(L) for decoupling thevoltage regulator 200 from the load resistive element R_(L). The loadcapacitive element C_(L) may be provided in an integrated circuit withthe voltage regulator 200, or may be provided external to such anintegrated circuit. A smaller load capacitive element C_(L) may beemployed with the voltage regulator according the invention thanrequired with prior art voltage regulators, and therefore may beintegrated with the voltage regulator where, in prior art voltageregulators, a discrete component was required.

The voltage regulator 200 of FIG. 7 is suitable for delivering apositive output voltage V_(OUT), for which the first input voltageV_(IN1) can be positive and the second input voltage V_(IN2) can bezero, for example a ground potential. FIG. 8 illustrates an embodimentof a voltage regulator 300 suitable for delivering a negative outputvoltage V_(OUT) in which the first input voltage V_(IN1) can be zero,for example a ground potential, and the second input voltage V_(IN2) canbe negative. The embodiment of FIG. 8 comprises the same elements as theembodiment of FIG. 7, namely the output stage 110, the feedback network120, first and second voltage-to-current converters 150, 155 and theprimary current mirror stage 130. Differences in the architecture andinterconnection of these elements is described below.

Referring to FIG. 8, the output transistor stage 110 has its firstterminal 112 coupled to the second input 106, its second terminal 114coupled to the output 104, and its control terminal coupled to thesecond current path 162. The output stage 110 comprises an n-channeloutput transistor MN which is an n-channel MOSFET in a common sourceconfiguration, having a source coupled to the first terminal 112, adrain coupled to the second terminal 114, and a gate coupled to thecontrol terminal 116. The feedback network 120 is coupled between theoutput 104 and the first input 102. The load resistive element R_(L) iscoupled between the output 104 and the first input 102. The optionalload capacitive element C_(L) is coupled in parallel with the loadresistive element R_(L).

The first transconductance amplifier T1 of the first voltage-to-currentconverter 150 in the embodiment of FIG. 8 has its first non-invertinginput 153 arranged to receive the reference voltage V_(FB) from thefeedback node 108. The first inverting input 152 of the firsttransconductance amplifier T1 is coupled to the first input 102 via thefirst current sensing resistor R_(S1), and its first output 154 coupledto a third current converter transistor MP3 for controlling theconductivity of the third current converter transistor MP3. The thirdcurrent converter transistor MP3 is coupled between the first currentpath 160 and the first current sensing resistor R_(S1). The firstvoltage-to-current converter 150 is arranged to receive the first inputvoltage V_(IN1) applied at the first input 102 by means of the firstconnection 168. The first connection 168 conveys the first current I1controlled by the first voltage-to-current converter 150. Therefore, thefirst current I1 passes through the third current converter transistorMP3, the first current sensing resistor R_(S1) and the first connection168.

Continuing to refer to FIG. 8, the second transconductance amplifier T2of the second voltage-to-current converter 155 has its secondnon-inverting input 157 arranged to receive the reference voltageV_(REF), its first inverting input 156 is coupled to the first input 102via the second current sensing resistor R_(S2), and its second output158 is coupled to a fourth current converter transistor MP4 forcontrolling the conductivity of the fourth current converter transistorMP4. The fourth current converter transistor MP4 is coupled between thesecond current path 162 and the second current sensing resistor R_(S2).The second voltage-to-current converter 155 is arranged to receive thefirst input voltage V_(IN) applied at the first input 102 by means ofthe second connection 170. The second connection 170 conveys the secondcurrent I2 controlled by the second voltage-to-current converter 155.Therefore, the second current I2 passes through the fourth currentconverter transistor MP4, the second current sensing resistor R_(S2) andthe second connection 170. As in all embodiments, the first and secondconnections 168, 170 are separate, that is they provide independentcurrent paths, enabling the voltage-to-current conversion performed bythe second voltage-to-current converter 155 to be independent of thevoltage-to-current conversion performed by the first voltage-to-currentconverter 150. Nevertheless, because changes to the first current I1resulting from changes in the feedback voltage V_(FB) are reflected inthe second current I2 by the primary current mirror stage 130, thecontrol of the second current I2 due to the reference voltage V_(REF)can be linearly superimposed on the changes in second current I2 due tothe changes in the feedback voltage V_(FB). The third and fourth currentconverter transistors MP3, MP4, are PMOS transistors in contrast to therespective NMOS first and second current converter transistors MN1, MN2in the embodiment of FIG. 7.

The primary current mirror stage 130 illustrated in FIG. 8 comprises athird current mirror transistor MN3 and a fourth current mirrortransistor MN4, these both being NMOS transistors. The third and fourthcurrent mirror transistors MN3, MN4 have their sources coupled to thesecond input 106 for receiving the second input voltage V_(IN2) andtheir gates coupled together, thereby establishing common operatingconditions for the third and fourth current mirror transistors MN3, MN4.The third current mirror transistor MN3 has its drain coupled to thefirst current path 160 for receiving the first current I1, and its draincoupled to its gate for controlling the gate of both the third andfourth current mirror transistors MN3, MN4 with a common voltage. Thefourth current mirror transistor MN4 has its drain coupled to the secondcurrent path 162 for receiving the second current I2 reflected from thefirst current I1. In particular, the first current I1 and the secondcurrent I2 both flow from, respectively, the first and secondvoltage-to-current converters 150, 155 to the primary current mirrorstage 130, rather than in the opposite direction as in the embodiment ofFIG. 7. For a current mirror ratio M of one, the third and fourthcurrent mirror transistors MN3, MN4 are of equal size, whereas for othervalues of the current mirror ratio, the third and fourth current mirrortransistors MN3, MN4 can be of different sizes. The control terminal 116of the output transistor stage 110 is coupled to the second current path162. In operation, under quiescent conditions, the reference voltageV_(REF) causes target values of the first and second currents I1, I2 tobe established in, respectively, the first and second current paths 160,162, and a target output voltage V_(OUT) to be established at the output104, with a corresponding target feedback voltage V_(FB). Any subsequentdeviation of the output voltage V_(OUT) from the target voltage value,due to variation in the resistance of the load resistive element R_(L)will result in a change to the feedback voltage V_(FB) and to the firstand second currents I1, I2, such that the voltage in the second currentpath 162 operates to control the output transistor stage 110 to causethe output voltage V_(OUT) to be restored to the target voltage value.

FIG. 9 illustrates another embodiment of a voltage regulator 400 whichis suitable for delivering a positive output voltage V_(OUT), althoughnot suitable for LDO operation. The first input voltage V_(IN1), whichis applied at the first input 102, can be positive and the second inputvoltage V_(IN2), which is applied at the second input 106 can be zero,for example a ground potential. Referring to FIG. 9, the outputtransistor stage 110 has its first terminal 112 coupled to the firstinput 102, its second terminal 114 coupled to the output 104, and itscontrol terminal 116 coupled to the second current path 162. The outputtransistor stage 110 comprises the n-channel output transistor MN in acommon drain configuration, having its drain coupled to the firstterminal 112, its source coupled to the second terminal 114, and itsgate coupled to the control terminal 116. Due to the use of the commondrain configuration, the voltage applied at the control terminal 116must exceed the output voltage V_(OUT) by at least the gate-sourcethreshold voltage of the n-channel output transistor MN, and thereforeLDO operation is not provided. The feedback network 120 is coupledbetween the output 104 and the second input 106. The load resistiveelement R_(L) is coupled between the output 104 and the second input102. The optional load capacitive element C_(L) is coupled in parallelwith the load resistive element R_(L).

The first transconductance amplifier T1 of the first voltage-to-currentconverter 150 in the embodiment of FIG. 9 has its first non-invertinginput 153 arranged to receive the reference voltage V_(REF), andtherefore for convenience is illustrated on the left of FIG. 9.Consequently, in FIG. 9 the first current path 160 is illustrated on theleft of the second current path 162. The first inverting input 152 ofthe first transconductance amplifier T1 is coupled to the second input106 via the first current sensing resistor R_(S1) and the firstconnection 168, and its first output 154 is coupled to the first currentconverter transistor MN1 for controlling the conductivity of the firstcurrent converter transistor MN1. The first current converter transistorMN1 is coupled between the first current path 160 and the first currentsensing resistor R_(S1). The first current I1 passes through the firstcurrent converter transistor MN1, the first current sensing resistorR_(S1) and the first connection 168.

Continuing to refer to FIG. 9, the second transconductance amplifier T2of the second voltage-to-current converter 155 has its secondnon-inverting input 157 arranged to receive the feedback voltage V_(FB)from the feedback node 108, its first inverting input 156 is coupled tothe second input 106 via the second current sensing resistor R_(S2) andthe second connection 170, and its second output 158 is coupled to thesecond current converter transistor MN2 for controlling the conductivityof the second current converter transistor MN2. The second currentconverter transistor MN2 is coupled between the second current path 162and the second current sensing resistor R_(S2). The second current I2passes through the second current converter transistor MN2, the secondcurrent sensing resistor R_(S2) and the second connection 170. The firstand second current converter transistors MN1, MN2, are NMOS transistors,as in the embodiment of FIG. 7.

The primary current mirror stage 130 illustrated in FIG. 9 is identicalto the primary current mirror stage 130 illustrated in, and describedwith reference to, FIG. 7, except that the positions of the first andsecond current mirror transistors MP1, MP2 are swapped to correspond tothe positions of the first and second current paths 160, 162. Inoperation, any deviation of the output voltage V_(OUT) from the targetvoltage value will result in a change to the feedback voltage V_(FB) andto the second current I2, such that the voltage in the second currentpath 162 operates to control the output transistor stage 110 to causethe output voltage V_(OUT) to be restored to the target voltage value.In addition, control exerted on the first current I1 by the firstvoltage-to-current converter 150 in response to the reference voltageV_(REF) is reflected to the second current I2 by the primary currentmirror stage 130, and contributes to establishing the target voltagevalue of the output voltage V_(OUT).

FIG. 10 illustrates another embodiment of a voltage regulator 500 whichis suitable for delivering a negative output voltage V_(OUT), althoughnot suitable for LDO operation. The first input voltage V_(IN1), whichis applied at the first input 102, can be zero, for example a groundpotential, and the second input voltage V_(IN2), which is applied at thesecond input 106 can be negative. Referring to FIG. 10, the outputtransistor stage 110 has its first terminal 112 coupled to the secondinput 106, its second terminal 114 coupled to the output 104, and itscontrol terminal 116 coupled to the second current path 162. The outputtransistor stage 110 comprises the p-channel output transistor MP in acommon drain configuration, having its drain coupled to the firstterminal 112, its source coupled to the second terminal 114, and itsgate coupled to the control terminal 116. Due to the use of the commondrain configuration, the voltage applied at the control terminal 116must be less than the output voltage V_(OUT) by at least the gate-sourcethreshold voltage of the output transistor MP, and therefore LDOoperation is not provided. The feedback network 120 is coupled betweenthe output 104 and the first input 102. The load resistive element R_(L)is coupled between the output 104 and the first input 102. The optionalload capacitive element C_(L) is coupled in parallel with the loadresistive element R_(L).

The first transconductance amplifier T1 of the first voltage-to-currentconverter 150 in the embodiment of FIG. 10 has its first non-invertinginput 153 arranged to receive the reference voltage V_(REF), andtherefore for convenience is illustrated on the left of FIG. 10.Consequently, in FIG. 10 the first current path 160 is illustrated onthe left of the second current path 162. The first inverting input 152of the first transconductance amplifier T1 is coupled to the first input102 via the first current sensing resistor R_(S1) and the firstconnection 168, and its first output 154 is coupled to the third currentconverter transistor MP3 for controlling the conductivity of the thirdcurrent converter transistor MP3. The third current converter transistorMP3 is coupled between the first current path 160 and the first currentsensing resistor R_(S1). The first current I1 passes through the thirdcurrent converter transistor MP3, the first current sensing resistorR_(S1) and the first connection 168.

Continuing to refer to FIG. 10, the second transconductance amplifier T2of the second voltage-to-current converter 155 has its secondnon-inverting input 157 arranged to receive the feedback voltage V_(FB)its second inverting input 156 coupled to the first input 102 via thesecond current sensing resistor R_(S2) and the second connection 170,and its second output 158 coupled to the fourth current convertertransistor MP4 for controlling the conductivity of the fourth currentconverter transistor MP4. The fourth current converter transistor MP4 iscoupled between the second current path 162 and the second currentsensing resistor R_(S2). The second current I2 passes through the fourthcurrent converter transistor MP4, the second current sensing resistorR_(S2) and the second connection 170. The third and fourth currentconverter transistors MP3, MP4, are PMOS transistors, as in theembodiment of FIG. 8.

The primary current mirror stage 130 illustrated in FIG. 10 is identicalto the primary current mirror stage 130 illustrated in, and describedwith reference to, FIG. 8, except that the positions of the third andfourth current mirror transistors MN3, MN4 are swapped to correspond tothe positions of the first and second current paths 160, 162. Inoperation, any deviation of the output voltage V_(OUT) from the targetvoltage value will result in a change to the feedback voltage V_(FB) andto the second current I2, such that the voltage in the second currentpath 162 operates to control the output transistor stage 110 to causethe output voltage V_(OUT) to be restored to the target voltage value.In addition, control exerted on the first current I1 by the firstvoltage-to-current converter 150 in response to the reference voltageV_(REF) is reflected to the second current I2 by the primary currentmirror stage 130, and contributes to establishing the target voltagevalue of the output voltage V_(OUT).

In order that the voltage regulator 100 has a fast operation, it isdesirable for the main feedback loop, formed by the output transistorstage 110, the feedback network 120, the first and secondvoltage-to-current converters 150, 155, the primary current mirror stage130 and the second current path 162, to have a high gain. The outputimpedance of the primary current mirror stage 130 contributes todetermining the open loop gain of the main feedback loop. If any errorsfrom the first and second voltage-to-current converters 150, 155 areneglected, then the open loop gain A of the main feedback loop can beapproximated at low frequencies by the expressionA=(gm_(MP)·R_(L))·(ro₁+ro₂)/(R_(S1)+R_(S2)) where gm_(MP) is thetransconductance of the output transistor stage 110, and in particularof the p-channel output transistor MP or the n-channel output transistorMN, R_(L) represents the resistance of a load resistive element R_(L)coupled to the output 104, ro₁ is the output resistance of the primarycurrent mirror stage 130 presented to the first current path 160, ro₂ isthe output resistance of the primary current mirror stage 130 presentedto the second current path 162, and R_(S1) and R_(S2) represent theresistance of, respectively, the first and second current senseresistors R_(S1), R_(S2).

The gain and bandwidth of the voltage regulator can be increased byadding a differential amplifier operating in parallel with the mainfeedback loop to provide an auxiliary feedback loop. Such embodimentsare illustrated in FIG. 11 for a voltage regulator 600 which is suitablefor delivering a positive output voltage V_(OUT), and in FIG. 12 for avoltage regulator 700 which is suitable for delivering a negative outputvoltage V_(OUT).

Referring to FIG. 11, the voltage regulator 600 comprises the sameelements as the voltage regulator 200 of FIG. 7, which therefore are notdescribed again except where additional features are included, and inaddition a differential amplifier 180 is coupled to the primary currentmirror stage 130 by means of a third current path 164 for conveying athird current I3 and is coupled to the primary current mirror stage 130by means of a fourth current path 166 for conveying a fourth current I4.In this illustrated arrangement, these couplings are via, respectively,a portion of the first and second current paths 160, 162. Therefore, inthis arrangement, a portion of the first current path 160 conveys notonly the first current I1 but also the third current I3, and a portionof the second current path 162 conveys not only the second current I2but also the fourth current I4. The primary current mirror stage 130delivers the sum of the first and third currents I1+I3 to the firstcurrent path 160, and the sum of the second and fourth currents I2+I4 tothe second current path 162. The primary current mirror stage 130controls the sum of the second and fourth currents I2+I4 dependent onthe sum of the first and third currents I1+I3 by reflecting the sum ofthe first and third currents I1+I3 such that the sum of the second andfourth currents I2+I4 is related to the sum of the first and thirdcurrents I1+I3 by the current mirror ratio M. The current mirror ratio Mmay have a value of one, in which case the sum of the first and thirdcurrents I1+I3 is equal to the sum of the second and fourth currentsI2+I4, or may be greater than one, in which case the sum of the secondand fourth currents I2+I4 exceeds the sum of the first and thirdcurrents I1+I3. Furthermore, the differential amplifier 180 is coupledto the feedback network 120 and is arranged to control the third currentI3 dependent on the feedback voltage V_(FB) and to control the fourthcurrent I4 dependent on the reference voltage V_(REF). In this way, inthe embodiment of FIG. 11, the primary current mirror stage 130 controlsboth the second current I2 and the fourth current I4 dependent on boththe first current I1 and the third current I3. In order to increase thestability and phase margin of the voltage regulator 600, it ispreferable for the third and fourth currents I3, I4 to be relativelysmall compared to, respectively, the first and second currents I1, I2,for example by a factor of at least ten.

In FIG. 11, the third current path 164 and the fourth current path 166are illustrated coupled to, respectively, the first and second currentpaths 160, 162 externally to the primary current mirror stage 130.However, equivalently, the third current path 164 and the fourth currentpath 166 can be coupled to, respectively, the first and second currentpaths 160, 162 internally to the primary current mirror stage 130.

In the embodiment illustrated in FIG. 11, the differential amplifier 180comprises a first differential amplifier transistor MN5 and a seconddifferential amplifier transistor MN6, these both being NMOStransistors. The first and second differential amplifier transistorsMN5, MN6 have their sources coupled to a current source 186 whichconveys the sum of the third and fourth currents I3+I4, and their drainscoupled to, respectively, the third current path 164 and the fourthcurrent path 166. The first differential amplifier transistor MN5 hasits gate coupled to the feedback node 108 for receiving the feedbackvoltage V_(FB), and the second differential amplifier transistor MN6 hasits gate coupled to the reference voltage V_(REF). Other embodiments ofthe differential amplifier 180 may alternatively be used.

Referring to FIG. 12, the voltage regulator 700 comprises the sameelements as the voltage regulator 300 of FIG. 8, which therefore are notdescribed again except where additional features are included, and inaddition the differential amplifier 180 is coupled to the primarycurrent mirror stage 130 by means of the third current path 164 forconveying the third current I3 and is coupled to the primary currentmirror stage 130 by means of the fourth current path 166 for conveyingthe fourth current I4. As in the embodiment of FIG. 11, a portion of thefirst current path 160 conveys not only the first current I1 but alsothe third current I3, and a portion of the second current path 162conveys not only the second current I2 but also the fourth current I4.The primary current mirror stage 130 receives the sum of the first andthird currents I1+I3 via the first current path 160, and the sum of thesecond and fourth currents I2+I4 via the second current path 162. Theprimary current mirror stage 130 controls the sum of the second andfourth currents I2+I4 dependent on the sum of the first and thirdcurrents I1+I3 by reflecting the sum of the first and third currentsI1+I3 such that the sum of the second and fourth currents I2+I4 isrelated to the sum of the first and third currents I1+I3 by the currentmirror ratio M. Again, the current mirror ratio M may have a value ofone, or may be greater than one, in the latter case the sum of thesecond and fourth currents I2+I4 exceeding the sum of the first andthird currents I1+I3. Furthermore, the differential amplifier 180 iscoupled to the feedback node 108 and is arranged to control the thirdcurrent I3 dependent on the feedback voltage V_(FB) and to control thefourth current I4 dependent on the reference voltage V_(REF). In thisway, in the embodiment of FIG. 12, the primary current mirror stage 130controls both the second current I2 and the fourth current I4 dependenton both the first current I1 and the third current I3. Again, in orderto increase the stability and phase margin of the voltage regulator 700,it is preferable for the third and fourth currents I3, I4 to berelatively small compared to, respectively, the first and secondcurrents I1, I2, for example by a factor of at least ten.

In FIG. 12, the third current path 164 and the fourth current path 166are illustrated coupled to, respectively, the first and second currentpaths 160, 162 externally to the primary current mirror stage 130.However, equivalently, the third current path 164 and the fourth currentpath 166 can be coupled to, respectively, the first and second currentpaths 160, 162 internally to the primary current mirror stage 130.

In the embodiment illustrated in FIG. 12, the differential amplifier 180comprises a third differential amplifier transistor MP5 and a fourthdifferential amplifier transistor MP6, these both being PMOStransistors. The third and fourth differential amplifier transistorsMP5, MP6 have their sources coupled to the current source 186 whichdelivers the sum of the third and fourth currents I3+I4, and theirdrains coupled to, respectively, the third current path 164 and thefourth current path 166. The third differential amplifier transistor MP5has its gate coupled to the feedback node 108 for receiving the feedbackvoltage V_(FB), and the second differential amplifier transistor MN6 hasits gate coupled to the reference voltage V_(REF). Other embodiments ofthe differential amplifier 180 may alternatively be used.

The gain and bandwidth of the voltage regulators 600, 700 of FIGS. 11and 12 can be increased by employing cascoded or wide-swing currentmirror circuitry in the primary current mirror stage 130 and couplingthe differential amplifier 180 to high impedance points of such currentmirror circuitry via the third and fourth current paths I3, I4. Anembodiment of the primary current mirror stage 130 employing suchwide-swing current mirror circuitry is illustrated in FIG. 13.

Referring to FIG. 13, the primary current mirror stage 130 comprises afifth current mirror transistor MP7 and a sixth current mirrortransistor MP8, these both being PMOS transistors. The fifth and sixthcurrent mirror transistors MP7, MP8 have their sources coupled to thefirst input voltage V_(IN1) and their gates coupled together, therebyestablishing common operating conditions for the fifth and sixth currentmirror transistors MP7, MP8. In addition, there is a seventh currentmirror transistor MP9 and an eighth current mirror transistor MP10,these also both being PMOS transistors. The seventh and eighth currentmirror transistors MP9, MP10 have their gates coupled together and to anon-illustrated bias voltage, their sources coupled to respective drainsof the fifth and sixth current mirror transistors MP7, MP8 and to thethird and fourth current paths 164, 166 respectively, and their drainsare coupled to the first and second current paths 160, 162 respectively.Therefore, the seventh and eighth current mirror transistors MP9, MP10conduct, respectively, the first and second current I1, I2, the fifthcurrent mirror transistor MP7 conducts the first and third currents I1,I3 in combination, and the sixth current mirror transistor MP8 conductsthe second and fourth currents I2, I4 in combination. If thedifferential amplifier 180 is balanced, then the third and fourthcurrents I3 and I4 are related by the current mirror ratio M and thebalance established in the bridge formed by the primary current mirrorstage 130, the first and second voltage-to-current converters 150, 155and the first and second current paths 160, 162 is maintained.

In a further embodiment, additional mirroring of currents may beemployed. Such an architecture enables a sliced based, that is, modular,approach to constructing a voltage regulator using a plurality of cellsof the same type. A single cell can be designed, and then repeated manytimes, according to the desired size of current to be delivered by thevoltage regulator.

FIG. 14 illustrates a voltage regulator 800 employing a single cellarchitecture. Referring to FIG. 14, the output transistor stage 110,which comprises the p-channel output transistor MP, has its firstterminal 112 coupled to the first input 102, its second terminal 114coupled to the output 104 and its control terminal 116 coupled to thesecond current path 162. The feedback network 120 is coupled between theoutput 104 and the second input 106. There is a secondary current mirrorstage 190 coupled to the first input 102 for receiving the first inputvoltage V_(IN1) and comprising a first secondary current mirror device192 and a second secondary current mirror device 194. The firstsecondary current mirror device 192 is coupled to the primary currentmirror stage 130 via the first current path 160 for conveying the firstcurrent I1, and is coupled to the first voltage-to-current converter 150via a third current path 196 for conveying a fifth current I5. Thesecond secondary current mirror device 194 is coupled to the primarycurrent mirror stage 130 via the second current path 162 for conveyingthe second current I2, and is coupled to the second voltage-to-currentconverter 155 via a fourth current path 198 for conveying a sixthcurrent I6. The first voltage-to-current converter 150 is coupled to thesecond input 106 via the first connection 168 for receiving the secondinput voltage V_(IN2) and for conveying the fifth current I5, andcontrols the fifth current I5 dependent on the reference voltageV_(REF). The second voltage-to-current converter 155 is coupled to thesecond input 106 via the second connection 170 for receiving the secondinput voltage V_(IN2) and for conveying the sixth current I6, and to thefeedback node 108 for receiving the feedback voltage V_(FB), andcontrols the sixth current I6 dependent on the feedback voltage V_(FB).As in all embodiments, the first and second connections 168, 170 areseparate, that is they provide independent current paths, enabling thevoltage-to-current conversion performed by the second voltage-to-currentconverter 155 to be independent of the voltage-to-current conversionperformed by the first voltage-to-current converter 150, but enablinglinear superposition in the second current I2 of the effects of thevoltage-to-current conversion performed by the first and secondvoltage-to-current converters 150, 155. The first voltage-to-currentconverter 150 and the second voltage-to-current converter 155 can have,for example, the internal architecture illustrated in FIG. 5.

In operation, the first secondary current mirror device 192 controls thefirst current I1 to be a reflection of the fifth current I5, the primarycurrent mirror stage 130 controls the second current to be a reflectionof the first current I1, and the second secondary current mirror device194 controls the second current I2 to be a reflection of the sixthcurrent I6. Therefore, changes in the sixth current I6 introduced by thesecond voltage-to-current converter 155 in response to changes in thefeedback voltage V_(FB) are reflected in the second current I2 by theseconds secondary current mirror device 194. Similarly, control of thefifth current I5 by the first voltage-to-current converter 150 inresponse to the reference voltage V_(REF) is reflected in the firstcurrent I1 by the first secondary current mirror device 192, andconsequently reflected in the second current I2 by the primary currentmirror stage 130 where they can be linearly superimposed on the changesin second current I2 due to the changes in the feedback voltage V_(FB).The first secondary current mirror device 192 and the second secondarycurrent mirror device 194 may operate with the same or different currentmirror ratios, which may be the same as, or different from, the currentmirror ratio M of the primary current mirror stage 130. Under quiescentconditions when the output voltage V_(OUT) is at the target voltagevalue, the current bridge formed by the primary current mirror stage130, the first and second current paths 160, 162 and the first andsecond voltage-to-current converters 150, 155 via the intermediary ofthe secondary current mirror stage 190, is in balance. As in the case ofthe other embodiments described, any deviation of the output voltageV_(OUT) from the target voltage value will result in a change to thefeedback voltage V_(FB) and to the first and second currents I1, I2,such that the voltage in the second current path 162 operates to controlthe output transistor stage 110 to cause the output voltage V_(OUT) tobe restored to the target voltage value. In FIG. 15, the embodiment ofFIG. 14 is extended to a voltage regulator 900 employing a three cellarchitecture, although other numbers of cells may be used. Referring toFIG. 15, the output transistor stage 110 comprises three sub-outputtransistors MPa, MPb, MPc each having a source coupled to the firstinput 102 via the first terminal 112 and each having a drain coupled tothe output 104 via the second terminal 114. A gate of each of the threesub-output transistors MPa, MPb, MPc is coupled to respective ones ofthree control sub-terminals 116 a, 116 b, 116 c which together form thecontrol terminal 116. In this way, the current delivered at the secondterminal 114 is sum of the three currents delivered to the secondterminal 114 by the three sub-output transistors MPa, MPb, MPc.

The first current path 160 comprises three first current sub-paths 160a, 160 b, 160 c for each conveying a proportion of the first current I1,and the second current path 162 comprises three second current sub-paths162 a, 162 b, 162 c for each conveying a proportion of the secondcurrent I2. Each of the three control sub-terminals 116 a, 116 b, 116 cis coupled to a different one of the three second current sub-paths 162a, 162 b, 162 c such that the conductivity of the respective sub-outputtransistors MPa, MPb, MPc between the first input 102 and the output 104is dependent on a voltage in the respective first current sub-paths 160a, 160 b, 160 c.

The primary current mirror stage 130 in the embodiment of FIG. 15.comprises three identical primary current mirror devices 130 a, 130 b,130 c each coupled to a respective one of the first current sub-paths160 a, 160 b, 160 c and a respective one of the second current sub-paths162 a, 162 b, 162 c, and each arranged to reflect the current in therespective one of the first current sub-paths 160 a, 160 b, 160 c in therespective one of the second current sub-paths 162 a, 162 b, 162 caccording to the current mirror ratio M.

The secondary current mirror stage 190 comprises three secondary currentmirror devices 192 a, 192 b, 192 c coupled to respective ones of thefirst current sub-paths 160 a, 160 b, 160 c. Three current mirrors areformed by each of the three secondary current mirror devices 192 a, 192b, 192 c being coupled to a common ninth current mirror transistor MP11which conducts the fifth current I5 current of the firstvoltage-to-current converter 150 and reflects that current to each ofthe first current sub-paths 160 a, 160 b, 160 c. Furthermore, thesecondary current mirror stage 190 comprises three further secondarycurrent mirror devices 194 a, 194 b, 194 c coupled to respective ones ofthe second current sub-paths 162 a, 162 b, 162 c. Three further currentmirrors are formed by each of the three further secondary current mirrordevices 194 a, 194 b, 194 c being coupled to a common tenth currentmirror transistor MP12 which conducts the sixth current I6 of the secondvoltage-to-current converter 155 and reflects that current to each ofthe second current sub-paths 162 a, 162 b, 162 c.

Each of the three cells may be constructed comprising one each of thesub-output transistors MPa, MPb, MPc, the primary current mirror devices130 a, 130 b, 130 c, the secondary current mirror devices 192 a, 192 b,192 c, the further secondary current mirror devices 194 a, 194 b, 194 c,the first current sub-paths 160 a, 160 b, 160 c and the second currentsub-paths 162 a, 162 b, 162 c. By employing identical cells andoperating conditions, the current in each cell is the same, and anarbitrary current can be delivered at the output 104 by employing anarbitrary number of the cells.

In the embodiment of FIG. 15, the feedback stage 120, the first andsecond voltage-to-current converters 150, 155 and the first and secondconnections 168, 170 are identical to the feedback stage 120, the firstand second voltage-to-current converters 150, 155 and the first andsecond connections 168, 170 in the embodiment of FIG. 14.

The voltage regulator 800 illustrated in FIG. 14 and the voltageregulator 900 illustrated in FIG. 15 are suitable for providing apositive output voltage V_(ouT). The secondary current mirror stage 190can also be employed in conjunction with voltage regulators forproviding a negative output voltage V_(OUT).

Referring to FIG. 16, an electronic apparatus 60 comprises a voltageregulator 62 in accordance with the invention and having the first input102 for the first input voltage V_(IN1) and the second input 106 for thesecond input voltage V_(IN2), which may be provided by, for example, abattery internal or external to the electronic device 60, and the output104 coupled to an application circuit 64 for delivering the outputvoltage V_(OUT) to the application circuit 64. The application circuit64 provides a load for the voltage regulator 62. The electronic device60 may be, for example, a mobile phone or a portable computer, or anintegrated circuit for use in such apparatus.

Additional embodiments of voltage regulators are described below whichillustrate other variations that fall within the scope of the invention,including the provision of plural independent output voltages fromdifferent source voltages and grounds. These additional embodiments willbe recognized as variations of the regulators 400, 500, 800, and 900described above in connection with FIGS. 9, 10, 14, and 15. Despite thevariations employed in each of the embodiments of the voltage regulator,according to the terminology used throughout this description and theaccompanying claims, for each additional embodiment, a primary currentmirror stage controls a plurality of second currents in a respectiveplurality of second current paths to be a reflection of a first currentI1 in a first current path 160. A plurality of control terminals ofrespective ones of a plurality of output transistor stages in eachadditional embodiment are coupled to respective ones of the secondcurrent paths conveying the plurality of second currents, and eachadditional embodiment includes a plurality of second V-I converters anda respective plurality of feedback networks.

In this way, several regulators on an integrated circuit chip can usethe same reference current source, which may be advantageously placednear the bandgap reference. An extra transistor in the primary currentmirror 130 shown in FIG. 9 provides a reference current for a respectivesecondary regulator, which is substantially just a copy of part of thesingle regulator. The modified primary current mirror with all extracurrents must be a central block, but the secondary regulator(s) can beanywhere on the chip and need only one respective conductor for thereference current from the modified primary current mirror. Thesecondary regulator(s) can have different (local) grounds and different(local) supply voltages, and the resistances in their respectivefeedback networks can be chosen to generate respective, differentregulated voltage(s).

FIG. 17 illustrates a voltage regulator 400-1 having the same generalarchitecture as the voltage regulator 400 illustrated in FIG. 9 butincluding a secondary regulator stage. Components in FIG. 17 that arethe same as components in FIG. 9 have the same reference numerals. Thesecondary regulator stage includes a secondary output stage 110-2, asecondary feedback network 120-2, and a secondary secondvoltage-to-current converter 155-2. It should be understood that FIG. 17depicts an arrangement having one secondary regulator stage configuredfor two regulated output voltages V_(OUT), V_(OUT)-2, but more than twooutput voltages can be provided by including additional secondaryregulator stages and additional transistors in the primary currentmirror 130.

In FIG. 17, a secondary first input voltage V_(IN1)-2 is applied at asecondary first input 102-2 and can be positive, and a secondary secondinput voltage V_(IN2)-2 is applied at a secondary second input 106-2 andcan be zero, for example a ground potential. The voltages V_(IN1),V_(IN1)-2 need not be the same, although those voltages need to be suchthat the working conditions of the transistors 110, 110-2 are correct toprovide the regulated voltages V_(OUT), V_(OUT)-2. This can be anadvantage if the secondary output stage 110-2 is supplied by a separateDC-DC converter or other source that has some tolerance against thesupply of the output stage 110. In addition, the “grounds” V_(IN2),V_(IN2)-2 can also be slightly different, e.g., 100 mV, which can helpto refer the secondary regulated voltage V_(OUT)-2 to the local groundvalue that can differ from the ground voltage on other points of a chipdue to currents or different ground domains, such as different pads to acommon external ground plane.

Because these are current mirrors, if the voltage at the sources of bothmirror transistors changes, the output current is still the same (or atleast nearly the same). It will be noted that the two transistors of onecurrent mirror (the ones with connected gates) must have the same sourcevoltage (i.e., the same supply or ground). In the voltage regulator400-1 depicted in FIG. 17, since V_(IN1)-2 is used only for transistor110-2, not much difference between V_(IN1) and V_(IN1)-2 can betolerated because it changes the voltage at secondary second currentpath 162-2 as well.

It is also believed to be important for the “ground” voltage of a V-Iconverter and its respective feedback network to be the same. Thus,V_(IN2) should be the same for converter 155 and network 120, and inFIG. 17, V_(IN2)-2 should be the same for the converters 155-2 andnetwork 120-2. Furthermore, resistors used in the second V-I converters155, 155-2 should have the substantially the same values, sizes,orientations, and surroundings because otherwise, the matching with thefirst V-I converter 150 can be bad and the regulator(s) output(s) canhave a tolerance against V_(REF).

As in FIG. 9, the secondary output transistor stage 110-2 has its firstterminal 112-2 coupled to the first input 102-2, its second terminal114-2 coupled to the output 104-2, and its control terminal 116-2coupled to the secondary second current path 162-2. The secondary outputtransistor stage 110-2 also comprises an n-channel output transistorMN-2 in a common drain configuration, having its drain coupled to thefirst terminal 112-2, its source coupled to the second terminal 114-2,and its gate coupled to the control terminal 116-2. Due to the use ofthe common drain configuration, the voltage applied at the controlterminal 116-2 must exceed the output voltage V_(ouT)-2 by at least thegate-source threshold voltage of the n-channel output transistor MN-2,else LDO operation is not provided. The secondary feedback network 120-2is coupled between the output 104-2 and the second input 106-2. Asecondary load resistive element R_(L)-2 is coupled between thesecondary output 104-2 and the secondary second input 106-2. An optionalsecondary load capacitive element C_(L)-2 is coupled in parallel withthe load resistive element R_(L)-2.

The secondary regulator includes a secondary second transconductanceamplifier T2-2 of a secondary second voltage-to-current converter 155-2,which has its second non-inverting input 157-2 arranged to receive asecondary feedback voltage V_(FB)-2 from a secondary feedback node 108-2of the secondary feedback network 120-2, its first inverting input 156-2coupled to the second input 106-2 via a second current sensing resistorR_(S2)-2 and a second connection 170-2, and its second output 158-2coupled to a secondary second current converter transistor MN2-2 forcontrolling the conductivity of the transistor MN2-2. The secondarysecond current converter transistor MN2-2 is coupled between thesecondary second current path 162-2 and the secondary second currentsensing resistor R_(S2)-2. The secondary second current I2-2 passesthrough the secondary second current converter transistor MN2-2, thesecondary second current sensing resistor R_(S2)-2 and the secondarysecond connection 170-2. The secondary second current convertertransistor MN2-2 is an NMOS transistor.

The primary current mirror stage 130-2 illustrated in FIG. 17 is amodification of the primary current mirror stage shown in FIG. 9. Asshown in FIG. 17, the primary current mirror stage 130-2 includesanother second current mirror transistor MP2-2 that is a PMOStransistor, and the transistor MP2-2 has its source coupled to the firstinput 102 for receiving the first input voltage V_(IN1) and its gatecoupled to the gate of the other second current mirror transistor MP2,thereby establishing a common operating condition. The secondary secondcurrent mirror transistor MP2-2 has its drain coupled to the secondarysecond current path 162-2 for delivering the secondary second currentI2-2 reflected from the first current I1. Other embodiments of theprimary current mirror stage 130-2 may alternatively be used.

In operation, any deviation of the secondary output voltage V_(OUT)-2from a target voltage value will result in a change to the secondaryfeedback voltage V_(FB)-2 and to the secondary second current I2-2, suchthat the voltage in the secondary second current path 162-2 operates tocontrol the secondary output transistor stage 110-2 to cause thesecondary output voltage V_(OUT)-2 to be restored to the target voltagevalue. In addition, control exerted on the first current I1 by the firstvoltage-to-current converter 150 in response to the reference voltageV_(REF) is reflected to the secondary second current I2-2 by the primarycurrent mirror stage 130-2, and contributes to establishing the targetvoltage value of the secondary output voltage V_(OUT)-2.

FIG. 18 illustrates a voltage regulator 500-1 having the same generalarchitecture as the voltage regulator 500 illustrated in FIG. 10 butincluding a secondary regulator stage. Components in FIG. 18 that arethe same as components in FIG. 10 have the same reference numerals. Thesecondary regulator stage includes a secondary output stage 110-2, asecondary feedback network 120-2, and a secondary secondvoltage-to-current converter 155-2. It should be understood that FIG. 18depicts an arrangement having one secondary regulator stage configuredfor two regulated output voltages V_(OUT), V_(OUT)-2, but more than twooutput voltages can be provided by including additional secondaryregulator stages and additional transistors in the primary currentmirror 130-2.

The secondary first input voltage V_(IN1)-2, which is applied at asecondary first input 102-2, can be zero, for example a groundpotential, and the secondary second input voltage V_(IN2)-2, which isapplied at a secondary second input 106-2 can be negative. Referring toFIG. 18, a secondary output transistor stage 110-2 has its firstterminal 112-2 coupled to the second input 106-2, its second terminal114-2 coupled to the output 104-2, and its control terminal 116-2coupled to a secondary second current path 162-2. The output transistorstage 110-2 comprises a p-channel output transistor MP in a common drainconfiguration, having its drain coupled to the first terminal 112-2, itssource coupled to the second terminal 114-2, and its gate coupled to thecontrol terminal 116-2. Due to the common drain configuration, thevoltage applied at the control terminal 116-2 must be less than thesecondary output voltage V_(OUT)-2 by at least the gate-source thresholdvoltage of the output transistor MP, and therefore LDO operation is notprovided.

The secondary feedback network 120-2 is coupled between the secondaryoutput 104-2 and the secondary first input 102-2. A secondary loadresistive element R_(L)-2 is coupled between the secondary output 104-2and the secondary first input 102-2. An optional secondary loadcapacitive element C_(L)-2 is coupled in parallel with the secondaryload resistive element.

A secondary second transconductance amplifier T2-2 of the secondarysecond voltage-to-current converter 155-2 has its second non-invertinginput 157-2 arranged to receive a secondary feedback voltage V_(FB)-2,its second inverting input 156-2 coupled to the secondary first input102-2 via a secondary second current sensing resistor R_(S2)-2 and asecondary second connection 170-2, and its second output 158-2 coupledto a secondary fourth current converter transistor MP4-2 for controllingthe conductivity of the secondary fourth current converter transistorMP4. The secondary fourth current converter transistor MP4 is coupledbetween a secondary second current path 162-2 and the secondary secondcurrent sensing resistor R_(S2)-2. A secondary second current I2-2passes through the secondary fourth current converter transistor MP4,second current sensing resistor R_(S2)-2, and secondary secondconnection 170-2. The third, fourth and secondary fourth currentconverter transistors MP3, MP4 and MP4-2 are PMOS transistors, as in theembodiment of FIG. 8.

The primary current mirror stage 130-2 illustrated in FIG. 18 is amodification of the primary current mirror stage 130 illustrated in, anddescribed with reference to, FIGS. 8 and 10. As in FIG. 10, thepositions of third and fourth current mirror transistors MN3, MN4 areswapped to correspond to the positions of first and second current paths160, 162. In addition as shown in FIG. 18, the primary current mirrorstage 130-2 includes a secondary fourth current mirror transistor MN4-2that has its source coupled to the second input 106 for receiving thesecond input voltage V_(IN2) and its gate coupled to the gate of thefourth current mirror transistor MN4, thereby establishing a commonoperating condition. The secondary fourth current mirror transistorMN4-2 has its drain coupled to the secondary second current path 162-2for delivering the secondary second current I2-2 reflected from thefirst current I1. Other embodiments of the primary current mirror stage130-2 may alternatively be used.

In operation, any deviation of the secondary output voltage V_(ouT)-2from a target voltage value will result in a change to the secondaryfeedback voltage V_(FB)-2 and to the secondary second current I2-2, suchthat the voltage in the secondary second current path 162-2 operates tocontrol the secondary output transistor stage 110-2 to cause thesecondary output voltage V_(OUT)-2 to be restored to the target voltagevalue. In addition, control exerted on the first current I1 by the firstvoltage-to-current converter 150 in response to the reference voltageV_(REF) is reflected to the secondary second current I2-2 by the primarycurrent mirror stage 130-2, and contributes to establishing the targetvoltage value of the secondary output voltage V_(OUT)-2.

FIG. 19 illustrates a voltage regulator 800-1 having the same generalsingle-cell architecture as the voltage regulator 800 illustrated inFIG. 14 but including a secondary regulator stage. Components in FIG. 19that are the same as components in FIG. 14 have the same referencenumerals. The secondary regulator stage includes a secondary outputstage 110-2, a secondary feedback network 120-2, and a secondary secondvoltage-to-current converter 155-2. It should be understood that FIG. 19depicts an arrangement having one secondary regulator stage configuredfor two regulated output voltages V_(OUT), V_(OUT)-2, but more than twooutput voltages can be provided by including additional secondaryregulator stages and additional transistors in the primary currentmirror 130-2. It should also be understood that one or more secondaryregulator stages can be added in a straightforward way based on FIG. 19to the voltage regulator 900 depicted in FIG. 15 because, as explainedabove, the three-cell architecture of FIG. 15 is an extension of thesingle-cell architecture of FIG. 14.

In FIG. 19, the secondary output transistor stage 110-2 comprises ap-channel output transistor MP having its first terminal 112-2 coupledto a secondary first input 102-2, its second terminal 114-2 coupled to asecondary output 104-2, and its control terminal 116-2 coupled to asecondary second current path 162-2. The secondary feedback network120-2 is coupled between the secondary output 104-2 and a secondarysecond input 106-2.

There is a secondary second current mirror stage coupled to thesecondary first input 102-2 for receiving a secondary first inputvoltage V_(IN1)-2 and comprising a secondary second current mirrordevice 194-2 that is coupled to the primary current mirror stage 130-2via the secondary second current path 162-2 for conveying a secondarysecond current I2-2, and is coupled to the secondary secondvoltage-to-current converter 155-2 via a secondary fourth current path198-2 for conveying a secondary sixth current I6-2. The secondary secondvoltage-to-current converter 155-2 is coupled to the secondary secondinput 106-2 via a secondary second connection 170-2 for receiving asecondary second input voltage V_(IN2)-2 and for conveying the secondarysixth current I6-2, and to the secondary feedback node 108-2 forreceiving the secondary feedback voltage V_(FB)-2. The secondary secondvoltage-to-current converter 155-2 controls the secondary sixth currentI6-2 dependent on the secondary feedback voltage V_(FB)-2 in allembodiments. The first and second connections 168, 170 and the secondarysecond connection 170-2 are separate, that is they provide independentcurrent paths, enabling the voltage-to-current conversion performed bythe secondary second voltage-to-current converter 155-2 to beindependent of the voltage-to-current conversion performed by theconverters 150, 155, but enabling linear superposition in the secondarysecond current I2-2 of the effects of the voltage-to-current conversionperformed by the first and secondary second voltage-to-currentconverters 150, 155-2. The first voltage-to-current converter 150 andthe secondary second voltage-to-current converter 155-2 can have, forexample, the internal architecture illustrated in FIG. 5.

In operation, the secondary second current mirror device 194-2 controlsthe secondary second current I2-2 to be a reflection of the secondarysixth current I6-2. Therefore, changes in the secondary sixth currentI6-2 introduced by the secondary second voltage-to-current converter155-2 in response to changes in the secondary feedback voltage V_(FB)-2are reflected in the secondary second current I2-2 by the secondarysecond current mirror device 194-2. Similarly, control of the fifthcurrent I5 by the first voltage-to-current converter 150 in response tothe reference voltage V_(REF) is reflected in the first current I1 andin the secondary second current I2-2 by the primary current mirror stage130-2, where they can be linearly superimposed on changes in thesecondary second current I2-2 due to changes in the secondary feedbackvoltage V_(FB)-2.

As in the embodiments described above, if the voltage at the sources ofboth mirror transistors changes, the output current is still the same(or at least nearly the same) due to the current mirrors. It will benoted that the two transistors of one current mirror (the ones withconnected gates) must have the same source voltage (i.e., the samesupply or ground). Because the V-I converters are connected to PMOSmirrors that in turn are driven by an NMOS mirror, both PMOS transistors194 can have a slightly different source voltage than the pair oftransistors 192. The voltage between the sources of the pair 192 and thesources of the pair 194 can differ, but not so much that the maximumvoltage at their drains comes above their rated value. Compared to FIG.17, more of a difference between V_(IN1) and V_(IN1)-2 can be toleratedbecause the sources of transistor pair 194-2 and transistor MP-2 areconnected to V_(IN1)-2. In this case, V_(IN1)-2 can be more than 100 mVdifferent from V_(IN1).

The first secondary current mirror device 192 and the secondary secondcurrent mirror device 194-2 can operate with the same or differentcurrent mirror ratios, which can be the same as, or different from, thecurrent mirror ratio of the primary current mirror stage 130-2. Underquiescent conditions when the secondary output voltage V_(OUT)-2 is at atarget voltage value, the current bridge formed by the primary currentmirror stage 130-2, the first and secondary second current paths 160,162-2, and the first and secondary second voltage-to-current converters150, 155-2 via the intermediary of the secondary second current mirrorstage 194-2, is in balance. As in the case of the other embodimentsdescribed, any deviation of the secondary output voltage V_(OUT)-2 fromthe target voltage value will result in a change to the secondaryfeedback voltage V_(FB)-2 and to the first and secondary second currentsI1, I2-2, such that the voltage in the secondary second current path162-2 operates to control the secondary output transistor stage 110-2 tocause the secondary output voltage V_(OUT)-2 to be restored to thetarget voltage value.

Other variations and modifications will be apparent to the skilledperson. Such variations and modifications may involve equivalent andother features which are already known and which may be used instead of,or in addition to, features described herein. Features that aredescribed in the context of separate embodiments may be provided incombination in a single embodiment. Conversely, features which aredescribed in the context of a single embodiment may also be providedseparately or in any suitable sub-combination.

It should be noted that the term “comprising” does not exclude otherelements or steps, the term “a” or “an” does not exclude a plurality, asingle feature may fulfil the functions of several features recited inthe claims and reference signs in the claims shall not be construed aslimiting the scope of the claims. It should also be noted that theFigures are not necessarily to scale; emphasis instead generally beingplaced upon illustrating the principles of the present invention.

What is claimed is:
 1. A voltage regulator, comprising: a first input for a first input voltage; a second input for a second input voltage lower than the first input voltage; an output for an output voltage; an output transistor stage having a first terminal coupled to a first one of the first and second inputs, a second terminal coupled to the output, and a control terminal for controlling the conductivity of the output transistor stage between the first terminal and the second terminal; a feedback network coupled to the output and a second one of the first and second inputs, being different from the first one of the first and second inputs, and arranged to produce at a feedback node a feedback voltage dependent on the output voltage; a first current path for conveying a first current and a second current path for conveying a second current; a primary current mirror stage coupled to the first current path and to the second current path and arranged to control the second current dependent on the first current; a first voltage-to-current converter coupled to the first current path and arranged to control the first current dependent on one of the feedback voltage and a reference voltage, and a second voltage-to-current converter coupled to the second current path and arranged to control the second current dependent on the other of the feedback voltage and the reference voltage, wherein the voltage-to-current conversion provided by the first voltage-to-current converter is independent of the voltage-to-current conversion provided by the second voltage-to-current converter; wherein the control terminal is coupled to the second current path for controlling the conductivity of the output transistor stage dependent on a voltage in the second current path indicative of a deviation of the second current from a target current value dependent on the reference voltage for thereby reducing a deviation of the output voltage from a target voltage value.
 2. The voltage regulator of claim 1, wherein: the first voltage-to-current converter comprises a first transconductance amplifier having a first transconductance amplifier first input coupled to the second one of the first and second inputs via a first current sensing resistive element, a first transconductance amplifier second input arranged to receive the one of the feedback voltage and the reference voltage, and a first transconductance amplifier output coupled to control the conductivity of a first current converter transistor dependent on a difference between a voltage at the first transconductance amplifier first input and a voltage at the first transconductance amplifier second input, wherein the first current converter transistor is arranged to control the first current in the first current path; and the second voltage-to-current converter comprises a second transconductance amplifier having a second transconductance amplifier first input coupled to the second one of the first and second inputs via a second current sensing resistive element, a second transconductance amplifier second input arranged to receive the other of the feedback voltage and the reference voltage, and a second transconductance amplifier output coupled to control the conductivity of a second current converter transistor dependent on a difference between a voltage at the second transconductance amplifier first input and a voltage at the second transconductance amplifier second input, wherein the second current converter transistor is arranged to control the second current in the second current path.
 3. The voltage regulator of claim 2, wherein: the one of the first and second inputs is the first input and the other of the first and second inputs is the second input; and the output transistor stage comprises an output transistor having a p-channel, a source coupled to the first terminal, a drain coupled to the second terminal and a gate coupled to the control terminal.
 4. The voltage regulator of claim 2, wherein: the one of the first and second inputs is the first input and the other of the first and second inputs is the second input; and the output transistor stage comprises an output transistor having an n-channel, a drain coupled to the first terminal, a source coupled to the second terminal and a gate coupled to the control terminal.
 5. The voltage regulator of claim 2, wherein: the one of the first and second inputs is the second input and the other of the first and second inputs is the first input; and the output transistor stage comprises an output transistor having an n-channel, a source coupled to the first terminal, a drain coupled to the second terminal and a gate coupled to the control terminal.
 6. The voltage regulator of claim 2, wherein: the one of the first and second inputs is the second input and the other of the first and second inputs is the first input; the output transistor stage comprises an output transistor having a p-channel, a drain coupled to the first terminal, a source coupled to the second terminal and a gate coupled to the control terminal.
 7. The voltage regulator of claim 3, wherein: the first and second current converter transistors each comprise an n-channel; the first transconductance amplifier first input and the second transconductance amplifier first input are inverting inputs; and the first transconductance amplifier second input and the second transconductance amplifier second input are non-inverting inputs.
 8. The voltage regulator of claim 5, wherein: the first and second current converter transistors each comprise a p-channel; the first transconductance amplifier first input and the second transconductance amplifier first input are inverting inputs; and the first transconductance amplifier second input and the second transconductance amplifier second input are non-inverting inputs.
 9. The voltage regulator of claim 2, wherein the first current sensing resistive element and the first current converter transistor are arranged in the first current path and the second current sensing resistive element and the second current converter transistor are arranged in the second current path.
 10. The voltage regulator of claim 2, comprising: a first secondary current mirror stage coupled between the first current path and the first voltage-to-current converter for controlling the first current dependent on a reflection of a current in the first voltage-to-current converter; and a second secondary current mirror stage coupled between the second current path and the second voltage-to-current converter for controlling the second current dependent on a reflection of a current in the second voltage-to-current converter.
 11. The voltage regulator of claim 10, wherein: the first current path comprises a plurality of first current sub-paths for each conveying a proportion of the first current; the second current path comprises a plurality of second current sub-paths for each conveying a proportion of the second current; the primary current mirror stage comprises a plurality of primary current mirror devices; the first secondary current mirror stage comprises a plurality of first secondary current mirror devices coupled to respective ones of the primary current mirror device via the respective first current sub-paths; the second secondary current mirror stage comprises a plurality of second secondary current mirror devices coupled to respective ones of the primary current mirror devices via the respective second current sub-paths; and the output transistor stage comprises a plurality of output transistors coupled between the first one of the first and second inputs and the output, wherein each of the output transistors is coupled to a different one of the second current sub-paths for controlling the conductivity of the respective output transistor between the first one of the first and second inputs and the output dependent on a voltage in the respective second current sub-path.
 12. The voltage regulator of claim 1, wherein the primary current mirror stage is arranged to control the second current to be equal to the first current.
 13. The voltage regulator of claim 1, wherein the primary current mirror stage is arranged to control the second current to be greater than the first current.
 14. The voltage regulator of claim 1, further comprising a differential amplifier stage coupled to the primary current mirror stage by a third current path for conveying a third current and by a fourth current path for conveying a fourth current, and coupled to the feedback network for receiving the feedback voltage; wherein the differential amplifier stage is arranged to control the third current dependent on the one of the feedback voltage and the reference voltage and to control the fourth current dependent on the other of the feedback voltage and the reference voltage; and the primary current mirror stage is arranged to control the fourth current dependent on the third current.
 15. The voltage regulator of claim 14, wherein the differential amplifier stage is arranged to control the third current to be smaller than the first current and the fourth current to be smaller than the second current.
 16. The voltage regulator of claim 1, further comprising a capacitive element coupled between the output and the feedback node.
 17. The voltage regulator of claim 1, comprising a capacitive element coupled between the output and one of the first and second inputs.
 18. The voltage regulator of claim 1, formed in an integrated circuit.
 19. An electronic apparatus, comprising the voltage regulator as claimed in claim
 1. 20. A method of regulating an output voltage, comprising: producing a feedback voltage dependent on the output voltage; controlling a first current in a first current path dependent on one of the feedback voltage and a reference voltage by a first voltage-to-current converter; controlling a second current in a second current path dependent on the first current by a primary current mirror stage and controlling the second current dependent on the other of the feedback voltage and the reference voltage by a second voltage-to-current converter, wherein the voltage-to-current conversion provided by the first voltage-to-current converter is independent of the voltage-to-current conversion provided by the second voltage-to-current converter; and reducing a deviation of the output voltage from a target voltage value by controlling the output voltage dependent on a voltage in the second current path indicative of a deviation of the second current from a target current value dependent on the reference voltage.
 21. A voltage regulator, comprising: a plurality of first inputs for a respective plurality of first input voltages; a plurality of second inputs for a respective plurality of second input voltages lower than the first input voltages; a plurality of outputs for respective ones of a plurality of output voltages; a plurality of output transistor stages, each output transistor stage having a first terminal coupled to a first one of either the plurality of first inputs or the plurality of second inputs, a second terminal coupled to an output, and a control terminal for controlling a conductivity of the output transistor stage between the first terminal and the second terminal; a plurality of feedback networks, each feedback network being coupled to a respective output and a second one of either the plurality of first inputs or the plurality of second inputs, the second one being one of a plurality of inputs different from the first one's plurality of inputs, and configured for producing at a respective feedback node a respective feedback voltage dependent on a respective output voltage; a first current path for conveying a first current, and a plurality of second current paths for conveying a respective plurality of second currents; a primary current mirror stage coupled to the first current path and to the plurality of second current paths and configured for controlling the plurality of second currents dependent on the first current; a first voltage-to-current converter coupled to the first current path and configured for controlling the first current dependent on a reference voltage; and a plurality of second voltage-to-current converters, each second voltage-to-current converter coupled to a respective one of the second current paths and configured for controlling a respective second current dependent on a respective feedback voltage; wherein voltage-to-current conversion provided by the first voltage-to-current converter is independent of voltage-to-current conversion provided by the second voltage-to-current converters; and a control terminal of each output transistor stage is coupled to a respective second current path for controlling conductivity of the output transistor stage dependent on a voltage in the respective second current path indicative of a deviation of the respective second current from a target current value dependent on the reference voltage, thereby reducing a deviation of the respective output voltage from a target voltage value.
 22. The voltage regulator of claim 21, wherein: the first voltage-to-current converter comprises a first transconductance amplifier having a first transconductance amplifier first input coupled to a first input or to a second input via a first current sensing resistive element, a first transconductance amplifier second input arranged to receive the reference voltage, and a first transconductance amplifier output coupled to control a conductivity of a first current converter transistor dependent on a difference between a voltage at the first transconductance amplifier first input and a voltage at the first transconductance amplifier second input, wherein the first current converter transistor is arranged to control the first current in the first current path; and each second voltage-to-current converter comprises a second transconductance amplifier having a second transconductance amplifier first input coupled to a respective first input or to a respective second input via a second current sensing resistive element, a second transconductance amplifier second input arranged to receive a respective feedback voltage, and a second transconductance amplifier output coupled to control a conductivity of a second current converter transistor dependent on a difference between a voltage at the second transconductance amplifier first input and a voltage at the second transconductance amplifier second input, wherein the second current converter transistor is arranged to control a respective second current in a respective second current path.
 23. The voltage regulator of claim 22, further comprising: a first secondary current mirror stage coupled between the first current path and the first voltage-to-current converter for controlling the first current dependent on a reflection of a current in the first voltage-to-current converter; and a plurality of second secondary current mirror stages, each second secondary current mirror stage being coupled between a respective second current path and the respective second voltage-to-current converter for controlling the respective second current dependent on a reflection of a current in the respective second voltage-to-current converter.
 24. The voltage regulator of claim 21, wherein at least two of the plurality of first input voltages differ from each other, or at least two of the plurality of second input voltages differ from each other. 